In an effort to satisfy consumer demand for faster data rates, while at the same time striving to use the radio frequency spectrum most efficiently, many modern wireless communications technologies employ non-constant-envelope modulation formats. For example, 802.11g (or “Wi-Fi”) wireless local area network (WLAN) technology employs orthogonal frequency-division multiplexing (OFDM), which is non-constant envelope modulation format. Third generation (3G) wideband code division multiple access (W-CDMA) cellular technology employs quadrature phase shift keying (QPSK), which is also a non-constant envelope modulation format. Other and future technologies, such as the fourth generation (4G) Long Term Evolution (LTE) cellular communications technology, also use and contemplate the use of non-constant-envelope modulation formats.
Non-constant-envelope modulation formats typically result in signals having a high peak-to-average (PAR) ratio. To avoid distortion of these signals as they are amplified for transmission, the radio frequency power amplifier (RFPA) of a traditional transmitter (e.g., a quadrature-modulator-based transmitter) must be implemented as a linear RFPA. However, because linear RFPAs are not very power efficient, the requirement of a linear RFPA results in a sacrifice of efficiency for linearity. This efficiency versus linearity trade-off is highly undesirable, particularly when the transmitter is employed in battery-powered applications such as in a wireless handset or a wireless network interface card of portable computer.
Not only are linear RFPAs inefficient, they are also usually the dominant consumer of power in a transmitter. For this reason, substantial efforts have been made to improve the efficiencies of RFPAs. One proven and commonly used approach is to employ an envelope modulator to dynamically control the power supplied to the RFPA. This “dynamic power control” approach is illustrated in FIG. 1. An envelope modulator 100 operates to modulate a direct current (DC) supply voltage Vsupply according to amplitude variations in an input envelope signal Venv to produce a dynamic power supply signal Vout, which is used to power the RFPA 102. By controlling the power supplied to the RFPA 102 so that it dynamically tracks the input envelope signal Venv, the efficiency of the RFPA 102 is improved.
In general, dynamic power control can be applied in either an envelope tracking (ET) system or an envelope elimination and restoration (EER) system. Operation of the EER system is similar to operation of the ET system, except that in the EER system the envelope information is removed before the signal is introduced to the RF input RFin of the RFPA 102. Removing the envelope information prior to amplification obviates the need to employ a linear RFPA, thereby circumventing the linearity versus efficiency trade-off that plagues more conventional communications transmitters. The RFPA 102 in the EER system is typically implemented as a Class D, E or F switch-mode type of RFPA. When configured in this manner, the previously-removed envelope information is restored at the output of the RFPA 102 by modulating the drain (or collector) of the RFPA 102 with the dynamic power supply signal Vout as the switch-mode RFPA 102 amplifies the constant-envelope signal.
The envelope modulator 100 in FIG. 1 can be implemented in various ways. One approach is to use a linear regulator, which can be implemented using an operational amplifier, as shown in FIG. 2. When configured as an envelope modulator, the linear regulator 200 provides a dynamic power supply signal Vout (i.e., an envelope modulated power supply signal Vout) that linearly tracks the amplitude variations of the input envelope signal Venv.
In addition to its linear response, one attractive property of the linear regulator 200 is that it can react quickly to sudden changes in the input envelope signal Venv. Consequently, when used to implement the envelope modulator 100 in FIG. 1, the RFPA 102 is able to operate over a wide dynamic range of output power. However, a significant drawback of the linear regulator 200 is that it is inefficient for input signal amplitudes that are lower than the magnitude of the DC supply voltage Vsupply. This inefficiency increases as the voltage difference between the input signal and DC supply voltage Vsupply widens.
A more efficient alternative to implementing the envelope modulator 100 is a power conversion device known as a switch-mode converter. FIG. 3 is a simplified drawing of a typical switch-mode converter 300. The switch mode converter 300 includes a comparator 302 and a buck converter 304. The buck converter 304 includes a transistor 306 configured to operate as a switch, an inductor 308, and a capacitor 310. The comparator 302 is configured to operate as a pulse-width modulator, generating a pulse-width modulated (PWM) signal having pulse-widths that vary depending on the amplitude of the input envelope signal Venv compared to the amplitude of a triangular reference signal. The PWM signal is coupled to the gate of the transistor 306, so that the transistor 306 turns on and off, alternately coupling and decoupling the inductor 308 to and from the DC supply voltage Vsupply. The inductor 308 and capacitor 310 operate as a low-pass filter, which filters the inductor current before it is transferred to the load 312. The resulting output voltage is an envelope modulated power supply signal Vout which tracks the amplitude variations of the input envelope signal Venv.
Although the switch-mode converter 300 in FIG. 3 is more efficient than the linear regulator 200, it has a couple of well-known drawbacks. First, the switching action of the transistor 306 generates switching noise, some of which is introduced to the RFPA supply input despite the presence of the inductor 308/capacitor 310 low-pass filter. This switching noise can make it difficult to comply with noise limitation requirements imposed by communications standards. Second, the switch-mode converter 300 is not operable over wide bandwidths. This is attributable to the large gate capacitance of the transistor 306 (typically 10-30 pF on an integrated circuit), which limits the switching speed of the switching transistor 306 to only about 5 MHz or so. Accurate envelope tracking requires a switching frequency of twenty to fifty times higher than the required signal envelope bandwidth. However, because the signal envelope bandwidth of wide bandwidth applications is often 1 MHz or higher, switch-mode converters are not well-suited for generating dynamic power supply signals in wideband applications.
Given the need for an envelope modulator that is both efficient and capable of operating over a wide bandwidth, various techniques have been proposed to exploit the most desirable properties of the linear regulator 200 and switch-mode converter 300 while at the same time avoiding their drawbacks. FIG. 4 is a drawing of an ET system 400 of one such approach. The ET system 400 comprises an envelope modulator 402 and an RFPA 404. The envelope modulator 402 includes a linear regulator 406 (similar to the linear regulator 200 shown and described above in connection with FIG. 2), a hysteresis comparator 408, and a switch-mode converter 410 (similar to the buck converter 304 of the switch-mode converter 300 shown and described above in connection with FIG. 3).
The switch-mode converter 410 operates to generally track the envelope of the input envelope signal Venv. The linear regulator 406 engages to compensate for the switch-mode converter's inability to track high-frequency content in the input envelope signal Venv, and to filter out switching noise generated in the switch-mode converter 410 by use of a feedback mechanism. The hysteresis comparator 408 reacts to voltage drops across the current sense resistor 414 that exceed predetermined upper and lower hysteresis voltage thresholds, by turning a switching transistor 412 of the switch-mode converter 410 on or off in manner that satisfies the current demand of the RFPA 404. The hysteresis voltage thresholds of the hysteresis comparator 408 are determined based on the desired combination of average switching frequency and signal fidelity.
To optimize the efficiency of the envelope modulator 402, the resistance of the current sense resistor 414 must be made small compared to the load resistance (i.e., the resistance of the RFPA 404 presented to the output of the envelope modulator 402). A small resistance is also required to avoid distorting the envelope modulator output voltage Vout caused by the output amplifier of the linear regulator 406 saturating. Absent a small resistance, distortion can only be avoided by limiting the maximum allowable amplitude of envelope modulator output voltage Vout. However, this results in degraded efficiency. So, for all these reasons, the resistance of the current sense resistor 414 must be small.
The envelope modulator 402 is usually formed in an integrated circuit (IC). A typical RFPA 404 presents a load of about five ohms. Accordingly, to optimize efficiency of the envelope modulator 402 and avoid exceeding the operational range of the linear regulator output amplifier, the resistance of the current sense resistor 414 on the IC must be on the order of only an ohm or less. Unfortunately, a resistance of this value, which is both accurate and reproducible, is very difficult to realize using standard semiconductor fabrication processes.